Tracking receiver

ABSTRACT

A tracking receiver for automatically detecting and phase locking a satellite beacon signal under fixed doppler and high rates of change of doppler. The receiver has a third-order tracking and acquisition loop that includes a cos-limited tangent phase detector which has a phase error range greater than + OR 120*.

Gurak et al.

[ Apr. 15, 1975 TRACKING RECEIVER 3.713.149 1/1973 Brurier et al. 343/9W' [75] Inventors: Richard J. Gurak, Summit; Basil C. 1742361 6/1973331/25 Thompson, Lake Hopatcong; Gerald Rabow, Nutley; John P. Filo, I IBasking Ridge, a" of Ni; Arnold Primary Exammer-Maynard R. W1lbur SeipelPlantation Fla Assistant ExaminerT. M. Bliim Attorney, Agent, or Firm,R,S. Sciascia; L. I. Shrago [73] Assignee: The United States of America as1 represented by the Secretary of the Navy, Washington, DC.

[22] Filed: Aug. 7, 1973 57 ABSTRACT [21] Appl. No.: 386,232

A tracking receiver for automatically detecting and 52 US. Cl. 343/117;343/9 331/25 Phase locking 8 Satellite beam" Signal under fixed [51 Int.Cl. 6615 5/02 PP and high rates of Change P The [58] Field of Search343/9, 117- 331/25 Ceiver has a third-Order tracking and acquisition 'Pthat includes a cos-limited tangent phase detector [56] References Citedwhich has a phase error range greater than $120.

UNITED STATES PATENTS 3,569,853 3/1971 Wolejsza, Jr 331/25 4 Claims, 5Drawing Figures 1 asses /Z /5 /8 /9 /IO /25 T-BGHz l 700mHz 2ND 70mHzI.F SIGNAL PHASE lNPU ,$$g FILTER SQUAR DET.

/ 7 630m Hz /l l 2&2: e 11 SmHz X2 0 REE PHASE SHIFT 2o SWEEP cmc. LOGICSTATUS f F (s) FREQ MULT. I 5111 HZ 5m Hz SUMMING LOOP 1310 -1410 vcxoNETWORK FILTER PRESET TRANSMITTER TRACKING RECEIVER The presentinvention relates generally to long-range. extra-terrestrial,communication systems. and more particularly, to a receiving arrangementfor tracking signals transmitted from an orbiting communicationsatellite or space vehicle.

There are numerous communication systems which have an orbitingsatellite or space vehicle as part of the communication link. The groundreceiving apparatus of these systems usually is designed with a trackingfeature that utilizes a phase lock loop to detect and lock on theradiated signal. In its fundamental form, this loop usually consists ofa phase comparator. a voltage control oscillator and a loop filter.

Associated with the data signals transmitted from such space stationsare doppler components due to the satellite motion and any other motionswhich may occur when the receiving apparatus is not stationary. Themagnitude of these doppler shifts and their rate of change are usuallyonly generally known. Consequently, the phase lock loop, for effectiveoperation, must be capable of quickly locking to a phase mismatchcondition as large as possible. This locking and its maintenance isfurther complicated because of the presence of noise peaks at the lowsignallevels encountered in satellite communication systems.

It is, accordingly, a primary object of the present invention to providea receiver for tracking a beacon signal originating from an orbitingsatellite.

Another object of the present invention is to provide a trackingreceiver for use in a satellite communication system which contains aphase lock feature.

Another object of the present invention is to provide a receivingarrangement wherein a beacon signal transmitted from a remote radiatingdevice is detected and phase locked to a locally generated frequencystandard.

Another object of the present invention is to provide a communicationreceiver which utilizes signal detect acquisition and trackingtechniques that involve a socalled cos-limited, tangent phase detectorand a thirdorder tracking and acquisition phase loop.

Another object of the present invention is to provide atracking receiverwhich operates in a wide frequency search until signal detection and,thereafter, operates at an acquisition frequency sweep to phase lock thedetected signal to a locally generated reference frequency standard.

These and other objects, advantages and novel features of the inventionwill become apparent from the following detailed description of theinvention when considered in conjunction with the accompanying drawingswherein:

FIG. 1 is a box diagram showing the general arrangement of the trackingreceiver;

FIG. 2 is a box diagram of the signal detecting portion of the receiver;

FIG. 3 is a box diagram of a so-called cos-limited Tanlock phasedetector;

FIG. 4 is a simplified showing of the phase lock loop; and

FIG. 5'is a schematic diagram of the circuit which forms the entire loopfilter function.

Referring now to FIG. 1 of the drawings, which shows the generalconfiguration of the tracking receiver of the present invention, it willbe seen that the beacon signal transmitted from the orbiting satellite,which in the LII present case is a carrier with a 800Hz bi-phasemodulation, is initially detected by a tracking antenna 1 which .isadapted to scan pre-selected sectors of space. The

signal so detected which may have a frequency in the range of 7.257.75gHz plus doppler effects is fed to a first down converter 2 whichhas as its other input a signal derived from a 5mHz voltage controloscillator 3. This locally generated reference signal is fed to afrequency multiplier 4 which multiplies it by a factor anywhere between],310 to 1.410 and provides a major portion of the feedback gain of thetracking loop. The output of the first down converter 2 is, therefore. asignal in the 700mHz range, whose frequency depends upon, of course, theexact multiplying factor selected in the frequency synthesizer 4 and anyassociated doppler shifts. Thereafter. this signal is fed to a seconddown converter 5 which has as its other input a 630mHz signal that isalso locally generated by multiplying a reference 5mHz signal fromsource 6 by I26 in circuit 7. The results of this processing is a mHzsignal input to the IF amplifier 8.

The reference standard for the overall receiving system, theSmI-Izsource 6, is an atomic clock whose output frequency is extremelystable. Source 6 controls a frequency multiplier 4 and insures theaccuracy of the frequency multiplying factors that are manuallyselected. Multiplier 4, thus, does not introduce any significantfrequency error or drift into the system.

After IF amplification, the signal is fed to an IF filter 9 and,thereafter, to a signal squarer 10. By subjecting the IF signal tosquaring, the bi-phase modulation on the carrier is removed. and, at thesame time, the frequency of the signal is doubled for the phasecomparison operation subsequently performed. Filter 9, in one particularmodification, had a l.8kHz band width. The amplified IF signal is alsofed to a noise squarer 11 and. thereafter, to a signal detector circuit60. The purpose of this signal processing is to provide, as will be seenhereinafter, a signal detect condition at a C/KT carrier to noise poweras low as 30db. This is achieved by comparing the (CNR), signal in anarrow 1.8kHz band width to the (CNRL, in a 36kHz band width.

The manner in which this is accomplished is shown in detail in FIG. 2where the output of signal squarer I0 is fed to a low pass filter 12having a band width of only 25I-Iz. Thereafter, the output of thisfilter is amplified in dc amplifier 13 to yield a voltage which may bedesignated -K(CNR) Likewise, the output from noise squarer 11 is fed toa low-pass filter 14 whose pass band is 50Hz, double that of filter 12.The output of this filter is likewise amplified in a dc amplifier 15 toyield the quantity K(CNR),, In this case, the dc amplifier is operatedin a non-inversion mode manner.

The outputs of the two dc amplifiers are applied to a pair of seriesresistors 16 and 17 of equal magnitude, and the signal appearing at thejuncture of these resistors is fed to a dc operational amplifier 18 atits negative input terminal. Also coupled to this amplifier is areference voltage 19 whose magnitude is pre-selected to act as thethreshold signal detecting level.

The action of summing resistors 16 and 17 is to produce a dc differencevoltage,

When this voltage exceeds reference voltage 19 then a signal appears inthe output of amplifier 18 indicating signal detection.

Normally. when the receiver is in its wide band frequency search mode,the voltage control oscillator 3 has a sawtooth wave form applied to itwhich is derived from sweep circuit 20 and coupled to its controlelement through a summing network 21. In this mode of operation, theVCXO, 3, is varied by iIOkHz as determined on the output side offrequency multiplier 4. The slope of this sawtooth accomplishes thisfrequency sweep in one second. However, whenever a signal appears in theoutput of amplifier 18, this mode of operation is discontinued and thecomplementary logic circuits associated with sweep circuit 20 maintain avoltage on oscillator 3 corresponding to the voltage which existedthereat at the time of signal detect. By so locking the oscillator, loopstress due to frequency offset is eliminated.

A predetermined time after signal detect, the acquisition frequencysweep is activated and superimposed on any existing signal detectvoltage. This waiting period is incorporated in the system to takeadvantage of any fortuitous locking which may occur as a result offavorable signal conditions within the loop and to enable the antennasto stop the spacial scan and lock on to the satellite. However, oncethis time interval expires, sawtooth circuit 20 applies an appropriatevoltage wave form to oscillator 3 which causes it to sweep iZkI-Iz at arate of i333Hz per second, again, as measured after multiplication infrequency multiplier 4. This sweep range encompasses the maximumfrequency offset of l.4kHz. This 1.4kHz value represents maximumfrequency offset that may be detected at high CNR levels. Onceacquisition is achieved, the frequency sweep is discontinued and memorycircuits associated with sweep circuit 20 permanently impress a finaldetect and acquisition sweep voltage on the oscillator 3 until trackingstatus is changed.

The output of signal squarer also provides the input to the phasedetector 25 in the loop circuit. As best seen in FIG. 3, this output isfed to a pair of phase detectors 30 and 31 of conventional design whichhave quadraturely-phased signals derived from the frequency standard 6as their other inputs. More specifically, the output from referencestandard 6 is doubled in frequency in circuit 33 and, thereafter, thelOmHz signal resulting therefrom is fed directly to phase detector 31and through a 90 phase shift 34 to phase detector 30. As is well known,the outputs of these detectors are cos 6,. and sin 6,. where 6,.represents the phase difference or error between the compared signals.

The tracking receiver utilizes a variation of the socalled Tanlock phasedetector. For purposes of description, this detector is called acos-limited tangent phase detector.

Phase detectors 30 and 31, as seen in FIG. 3, form part of thiscos-limited tanlock detector. The output of these detectors 6,., asmentioned hereinbefore, correspond to the sin and cos of the phasedifference between 6, and 6 the compared signals. In the tanlockdetector, these signals feed the numerator and denominator inputs of ananalog multiplier divider whose output is in the form of l K) sin6,.I+F(K)cos6,. (H

K in the above equation is a parameter less than I which may be adjustedas desired. If K is set to equal 0, the equation becomes theconventional sinusoidal phase lock operation, and as the value of K isincreased, both the range and the linearity of the characteristicincrease. As a value of K equals l, the equation goes to i infinity forvalues of 6,. equal It would be pointed out that the name tanlock isderived from similarity between the above function and the trigometricidentity.

sin X The term cos-limited tangent phase detector is used to representthe fact that the term K cos 6, is limited so as to prevent thedenominator of the output function from reaching 0, a condition whichmay occur due to noise peaks at low CNR levels. This provisioneliminates serious dc offset errors during signal detect andacquisition.

Referring again to FIG. 3, the output of phase detector 30, cos 6,., isfed to a dc amplifier 40 having a first feedback path through resistor41 and a second feed back path through back-to-back Zener diodes 42 and43. The purpose of these series diodes is to effectively clamp theoutput signal at symmetrical positive and negative voltage levelscorresponding to the combined forward and reverse breakdown voltages ofthe two diodes. When these diodes break down, it will be appreciated,they effectively shunt feedback resistor 41 and reduce the gain of theamplifier.

The output of dc amplifier 40 is made to correspond to the quantity F(K)cos 6,, with the above clamping preventing F(K) from reaching 1.

The output from phase detector 31 which corresponds to sin 6,, islikewise fed to an operational amplifier 44 having a feedback resistor45, and this amplifier is adjusted to yield an output signalcorresponding to 1 K) sin 6, I

The signal appearing in the output circuit of amplifier 40 is coupled toa summing network 46 so as to have added to it a reference voltagecorresponding to +1 and.

the signal resulting therefrom which corresponds to l F(K) cos 6,. issupplied to an analog multiplier 47.

In a similar manner, the signal appearing in the output of amplifier 44is supplied to a summing network 48 so as to have added to it a signalcorresponding to the output signal of analog multiplier 47. The resultsof this addition is fed to a dc amplifier 49 which has a high gaincharacteristic. This amplifier provides the output signal from the phasedetector 25 as shown in FIG. 1. Additionally, this output signal E" l+F(K) c056,.

is also fed back to serve as the other input for analog multiplier 47.

In the usual tanlock circuit, the sin and cos error signals derived fromthe conventional phase detectors 30 and 31 are treated differently, thatis, these signals have a reference voltage added to the cos signal whilethe sin signal is multiplied by a reference voltage. The signalsresulting from this processing are divided to yield the above outputsignal.

In the system of the present invention, the same results are achieved byutilizing the summing networks 46 and 48, the high gain amplifier 49 andthe analog multiplier 47. It can be mathematically shown that when thegain of the amplifier49 is sufficiently high, its output will, for allpractical purposes, be equal to the input to summing network 48 dividedby the output from summing network 46.

This output voltage, E is fed to loop filter 50 and then to the voltagecontrol oscillator 3 to close the loop. In its basic form as shown inFIG. 4, this thirdorder loop has a high-gain, narrow band width. Itshould be appreciated that the arrangement shown in this FIG. 4 has beensimplified to facilitate a better understanding of the operation andperformance of the loop filter module 50 and that, for example, mixer 61represented as a component of this loop in actual practice comprisesthose circuits and signal sources of FIG. 1 which are responsible fordeveloping the IF signal which is present in the loop circuit. Itachieves approximately db additional loop gain at the highestanticipated rate of change of doppler frequency of l rad/sec compared toits second-order counterpart. It also enhances acquisition probabilityby reducing the acceleration phase errors by a factor of 10 to l atacquisition. The exact enhance mechanism is one of reducing theprobability of unlock once the loop has acquired the signal.

The transfer function of loop filter 50 is 1 +m) (1 +35 nn-(l+sn)(l+.s'1;,) (3) where s is a Laplace complex variable and 1,, r Tand 1 are time constants of the loop filter in seconds. The looptransfer function K F(s) where K 180,000 rad/sec This function containsa third-power Laplace complex variable in its denominator, and it is thepresence of this term which classifies the loop as a third-order loop.In a preferred modification of the invention, 1' was 2.86, 1' 0.0125, 11.2 and r 0.11. With this transfer function, the following trackingsystem constants were obtained:

Damping factor g 0.5

Single sided loop band width B 40Hz Loop natural resonant frequency W 80rad/sec More specifically, loop filter 50, as shown in FIG. 5, includesa first operational amplifier 51 operating in the non-inverting mode anda second operational amplifier 52 operating in the inverting mode.Amplifier 51 has an impedance Z, associated with its feedback circuitwhich is made up of resistor R in parallel with the v series combinationof resistor R and capacitor C. This circuitis connectedbetween theoutput and the negative input terminal, .which is grounded throughresistor R The input signal to this amplifier, which corresponds to"( lK) sin 0,.,;is obtained from phase detector 31. In a-preferredembodiment of the invention. R 1.000 ohms, R;= 5 lkohms, R 5. lkohms, C=23uf. For these parametersi it can be shown that nit-R..."

"u'hieh is the first term of H) in equation 13) The output fromamplifier 51 is fed to a divider circuit which is transparent to theloop filter function. This divider circuit, which is illustrated in aslightly different form in FIG. 3, has its dc amplifier 49 inserted inthis portion of the loop circuit as a convenient way of introducing intothe filtering system the signal 1 F(K) cos 0,, which is the denominatorof equation l defining the output of the tanlock phase detector.

The output of the dc amplifier 49 is supplied to the second operationalamplifier 52. Associated with this second operational amplifier is asimilar feedback impedance Z consisting again of R, in parallel withresistor R and capacitor C in series. Thus, 1', and 1,, C (R -l- R and 1and r CR. In the preferred embodiment of this amplifier R, kohms, R382kohms. R 1.69kohms, C= 7.5uf. v y

Utilizing the same relationships as s pecifiedin connection withoperational amplifier 51,. it ean be shown E R, *l+ I+j().llw

0.1 Is v E R, I L25 l+j L2! which is the second term of F(.\') inequation (3) With the receiving system disclosed, signal acquisitionexceeding probability may be achieved in the presence of an accelerationSOOHz/sec sinusoidal rate of change of doppler, tIOkHz doppler shift anda C/KT as low as 36db. This 90% probability is a result of theself-adaptive high gain, narrow band width, stable third-order loop andthe cos-limited tangent phase detector operating in the sawtooth sweepdetection and acquisition modes. It would be pointed out the tanlockphase detector has a phase lock range greater than il20 as compared tothe i907: of conventional circuits. It also has a more linear andgreater dynamic output capability.

It would be pointed out that tracking error information for pointing theantenna at the satellite may be obtained from the receivers automaticgain control circuit. Also, negative doppler correction information forthe transmitter loop may be obtained from the tracking loop to correctfor frequency errors caused when, for example, the receiver is part of ashipboard system. The SmHz VCXO contains positive doppler information,that is, the changes in frequency of this oscillator due to doppler arein the same direction of the doppler shifts themselves. A dopplercorrection circuit, now shown, inverts this doppler to a negativedoppler and, thereafter, a subsequent frequency multiplication in a 1310to a 1410 multiplier prepares this signal for a mixing operation toproduce a 7 to 8gHz signal for transmission back up to the satellite,

What is claimed is:

1. In a receiver for tracking a signal radiated from a remote source, atracking and acquisition loop comprising:

a voltage controlled local oscillator;

means for multiplying the frequency from the local oscillator by apre-selected number;

means for mixing the signal from said remote source with said multipliedsignal so as to produce an IF signal;

a cos-limited tangent phase detector having two pairs of input circuitsand an output circuit,

said coslimited tangent phase detector being controlled such that theoutput signal therefrom is prevented from increasing in magnitude beyonda predetermined level;

means for producing a pair of quadraturely phased reference signals atthe IF frequency;

means for feeding said IF signal and one of said reference signals toone pair of input circuits and for feeding said IF signal and the otherof said IF signals to the other pair of input circuits of saidcoslimited tangent phase detector;

a filter; and

means for feeding the output signal developed in the output circuit ofsaid cos-limited tangent phase detector to the input of said filter andfor feeding the output of said filter to the control element of saidvoltage controlled local oscillator.

2. ln an arrangement as defined in claim 1 wherein said filter has atransfer function such that the loop in which it is disposed whichincludes said local oscillator, said means for multiplying the frequencythereof and said cos-limited tangent phase detector is a third-orderloop.

3. ln a receiver for tracking a signal radiated from a remote source andfor locking on to said signal a phaselock loop comprising, incombination a voltage controlled local oscillator; means for multiplyingthe frequency of the signal produced by said local oscillator by apreselected number; means for mixing the signal radiated by said remotesource with said multiplied signal so as to produce an IF signal; acos-limited tangent phase detector; means for generating a pair ofquadraturely phased reference signals at the IF frequency; means forfeeding said quadraturely phased reference signals and said IF signal tosaid cos-limited tangent phase detector, whereby an output signalgenerally corresponding to (l K) sin0 /l F(K) cosl9 is produced where Kis a constant and 0,. is the phase error between said reference signaland said IF signal; means for controlling the operation of saidcoslimited tangent phase detector such that the quantity l F(K) c050 isprevented from equaling 0; a filter; and means for feeding the outputsignal from said coslimited tangent phase detector to the input of saidfilter and for coupling the output thereof to the control element ofsaid local oscillator, said filter, said local oscillator and saidcos-limited tangent phase detector being part of a third order loop. 4.In an arrangement as defined in claim 3 wherein said filter has atransfer function (l+s'r )(l+s1 (1+.w )(1+.

where 7,, 1 1- 1 are different time constants and s is the Laplacianoperator.

1. In a receiver for tracking a signal radiated from a remote source, a tracking and acquisition loop comprising: a voltage controlled local oscillator; means for multiplying the frequency from the local oscillator by a pre-selected number; means for mixing the signal from said remote source with said multiplied signal so as to produce an IF signal; a cos-limited tangent phase detector having two pairs of input circuits and an output circuit, said cos-limited tangent phase detector being controlled such that the output signal therefrom is prevented from increasing in magnitude beyond a predetermined level; means for producing a pair of quadraturely phased reference signals at the IF frequency; means for feeding said IF signal and one of said reference signals to one pair of input circuits and for feeding said IF signal and the other of said IF signals to the other pair of input circuits of said cos-limited tangent phase detector; a filter; and means for feeding the output signal developed in the output circuit of said cos-limited tangent phase detector to the input of said filter and for feeding the output of said filter to the control element of said voltage controlled local oscillator.
 2. In an arrangement as defined in claim 1 wherein said filter has a transfer function such that the loop in which it is disposed which includes said local oscillator, said means for multiplying the frequency thereof and said cos-limited tangent phase detector is a third-order loop.
 3. In a receiver for tracking a signal radiated from a remote source and for locking on to said signal a phase-lock loop comprising, in combination a voltage controlled local oscillator; means for multiplying the frequency of the signal produced by said local oscillator by a preselected number; means for mixing the signal radiated by said remote source with said multiplied signal so as to produce an IF signal; a cos-limited tangent phase detector; means for generating a pair of quadraturely phased reference signals at the IF frequency; means for feeding said quadraturely phased reference signals and said IF signal to said cos-limited tangent phase detector, whereby an output signal generally corresponding to (1 + K) sin theta e/1 + F(K) cos theta e is produced where K is a constant and theta e is the phase error between said reference signal and said IF signal; means for controlling the operation of said cos-limited tangent phase detector such that the quantity 1 + F(K) cos theta e is prevented from equaling 0; a filter; and means for feeding the output signal from said cos-limited tangent phase detector to the input of said filter and for coupling the output thereof to the control element of said local oscillator, said filter, said local oscillator and said cos-limited tangent phase detector being part of a third order loop.
 4. In an arrangement as defined in claim 3 wherein said filter has a transfer function 